Energy Extraction From The Parasitic Elements In Power Converters

ABSTRACT

A switching power conversion apparatus for converting power from an input voltage source to a load includes first and second switches connected to a switching node. An inductive element has a magnetizing current connected to the node, and the inductive element is connected to deliver energy via the first and second switches from the input voltage to the load during a succession of power conversion cycles. A capacitance connected to the node resonates with the inductive element to cause parasitic oscillation. A clamp subcircuit across the inductive element contains an auxiliary switch to trap energy and prevent parasitic oscillation, wherein the auxiliary switch is complementary to the first switch. A controlled voltage source injects energy in the inductive element, when the auxiliary switch turns off to discharge the parasitic capacitance by using trapped energy in the inductive element in addition to injected energy from the controlled voltage source.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation in part of and claims the benefit ofprior U.S. patent application Ser. No. 18/233,315 filed Aug. 12, 2023,is a continuation in part of and claims the benefit of prior U.S. patentapplication Ser. No. 18/199,959 filed May 21, 2023, and claims thebenefit of prior U.S. Provisional Application No. 63/377,229 filed Sep.27, 2022, and of prior U.S. Provisional Application No. 63/578,130 filedAug. 22, 2023.

FIELD

The present specification pertains to electronic devices employing apower converter configured around the topologies used in powerconverters.

BACKGROUND

All the topologies utilized in power conversion, naming just several,buck, boost, flyback, half bridge, half bridge derivative, full bridgeand full bridge derivative, contain switching devices connected toswitching nodes. In FIG. 1 is presented such a switching deviceconnected to a switching node. A capacitance C, 120 is connected to theswitching node, regardless of the type of the switching device. To theswitching node other devices are connected such additional switchingdevices such as Sn,130 and inductive elements such as Ln,140. Thespecific inductive element Ln has two terminations, terminal W1, 4100 ofthe inductive elements connected to the switching node, and the secondterminal of said inductor element, labeled, W2, 4110.

During the time all the switching devices connected to the switchingnode are off, the inductive elements connected to the switching nodewhich form a resonant circuit with the parasitic capacitance C,120 andparasitic oscillations do occur in the switching node. This period isreferred to as “dead time”. The oscillations which occur during the deadtime are named, “parasitic oscillations” and the energy contained in the“parasitic oscillations energy” is named the “parasitic energy”.

This specification will present a solution of extracting the “parasiticenergy” which is the energy contained in the resonant elements connectedto the switching node and as a result eliminating these oscillations. Inthe prior art a significant portion of this energy is dissipated due toconduction and inductive element magnetic core loss. In addition to thatthe “parasitic oscillations” will inject noise into the componentsplaced near the resonant elements. This specification presents solutionsin harvesting the energy of the “parasitic oscillations” and use thatenergy for improving the power conversion efficiency, such as creatingzero voltage switching conditions on the switching elements, this beingone of the embodiments of this specification. One Prior Art solution toeliminate the oscillation in the switching nodes is presented in U.S.Pat. No. 6,522,108B2. In this specification Prager presents aunidirectional switch which is placed across the inductive elementconnected to the switching node as is depicted in FIG. 2A. In FIG. 2Bare depicted the key waveforms of circuit from FIG. 2A.

The Prager′ patent presents a solution designed to eliminate the ringingbut does not offer a solution in using this energy to obtain zerovoltage switching conditions for the switching elements.

In FIG. 3A is presented a boost topology and in FIG. 3B are presentedthe key waveforms of a boost converter. The boost topology contains aninductive element L, 220, a switching element, M1,210, across whichthere is a parasitic capacitance Cp,300, parasitic capacitance being thesummation of the parasitic capacitance of the semiconductor, device usedas a switch, together with the parasitic capacitance reflected from theinductor element L and the rectifier means Do, 240. The boost topologycontains further a rectifier means, Do, and the output load which isformed by an output capacitor Co, 250 and an output loan R_(load), 330.

In between t0 to t1, the main switch M1, is turned on by the controlsignal VcM1, 270, and during this time the current through L,220, buildsup reaching a peak current Ipk, 360. During this time interval, energyis accumulated in the inductive element L.

At t1, the switch M1, 210, is turned off and the energy stored inbetween t0 to t1 is delivered to the “output load” formed by Co, 250 andthe load, R_(load), 330. As the energy is transferred to the outputload, the current through the inductive element is decreasing with arate given by the difference (Vo-Vin) and the inductance of L, 220. Att2, the current through Do, 240 will reach zero, which means that allthe energy stored in L, 220, is delivered to the output load.

In between t2 to t3, referred also as the dead time, “parasiticoscillations” will occur in the switching node, A, 280, depicted byV(M1) waveforms. The oscillations do have peaks, Vpk, 340 and valleys,Vv,350. These oscillations do reflect also in the current through L, 220which is depicted by the I_(L) waveforms, 290. The “parasiticoscillations” are caused by the resonant circuit formed by the inductiveelement L, 220 and the parasitic capacitance Cp, 300. Said resonantcircuit is energized by the energy defined by the formula presented inFIG. 3C.

In the case wherein the boost converter is used in Power FactorCorrection (PFC) application, the Wp can be quite significative at lowAC input voltage, in PFC application wherein Vin=127Vdc (90 Vac) and400V output. For a parasitic capacitance Cp=100 pF the Wp energy is 7.19uJ. For an operation frequency of 150 Khz the power dissipation in theparasitic ringing, if dissipated and not extracted, can reach 1 W.

This specification presents a method of extracting this energy and usethat energy for other purposes. One of the preferred embodiments isusing this energy to obtain zero voltage switching conditions for themain switch, M1, 210.

In the prior art presented in U.S. Pat. No. 6,522,108B2, the parasiticoscillations are eliminating by creating a short circuit across theinductive element and store that energy through the magnetizing currentof the inductive element. In FIG. 2A the short is created by the switchRs,30 and the diode 32. Preserving “parasitic oscillation” by shortingan inductive element has the disadvantage of dissipation of some of theenergy through the conduction in the inductive element and theunidirectional switch formed by Rs,30 and diode 32. For a larger “deadtime”, defined by the time difference t3−t2, this prior art methoddepicted in FIGS. 2A and 2B is not efficient and a good portion of theenergy would be dissipated in conduction. In this U.S. Pat. No.6,522,108B2 prior art the “parasitic oscillation” energy is not used forother purposes such as obtaining zero voltage switching conditions forthe switching elements. This drawback is eliminated by Jitaru in U.S.Pat. No. 11,152,847, wherein additional energy is injected to achievezero voltage switching on the switching elements, this patentapplication is a continuation in part. In Jitaru patent U.S. Pat. No.11,152,847 zero voltage switching on the switching elements is obtainedbut it is not the most efficient solutions especially for a larger “deadtime”.

BRIEF DESCRIPTION OF THE DRAWINGS

Referring to the drawings:

FIG. 1A depicts a switch node which is the building block of a powerconverter.

FIG. 1B depicts an intelligent clamp circuit.

FIG. 1C depicts the current injection module.

FIG. 2A depicts a prior art circuit.

FIG. 2B depicts the key waveforms of the circuit depicted in FIG. 2A.

FIG. 3A depicts a conventional boost topology.

FIG. 3B depicts the key waveforms from the circuit presented in FIG. 3A

FIG. 3C presents the formula for the energy contained in the parasiticoscillation of a boost topology.

FIG. 3D depicts a potential implementation of the circuit presented inFIG. 1B.

FIG. 3E depicts another potential implementation of the circuitpresented in FIG. 1B.

FIG. 3F depicts another potential implementation of the circuitpresented in FIG. 1B.

FIG. 4A depicts a buck topology using the clamp from FIG. 1B.

FIG. 4B depicts the key waveforms of the circuit from FIG. 4A.

FIG. 4C depicts the key waveforms of the circuit from FIG. 4A withintelligent control of Vvinjc.

FIG. 5A depicts the conventional buck topology.

FIG. 5B depicts the key waveforms of the circuit presented in FIG. 5A.

FIG. 5C presents the formula for energy contained in the parasiticoscillations.

FIG. 5D presents the buck topology with the intelligent clamp circuitfrom FIG. 1B.

FIG. 5E presents the key waveforms for the circuit depicted in FIG. 5D.

FIG. 6A presents the flyback topology.

FIG. 6B presents the key waveforms of the circuit from FIG. 6A.

FIG. 6C presents the formula calculating the energy in the parasiticoscillation of the flyback topology.

FIG. 7A presents the flyback topology using the intelligent clampcircuit placed on an auxiliary winding.

FIG. 7B presents the key waveforms of the circuit from FIG. 7A.

FIG. 8A presents the flyback topology using the intelligent clampcircuit and current injection module placed on auxiliary windings.

FIG. 8B presents the key waveforms of the circuit from FIG. 8A.

FIG. 9A presents the two transistors forward topology.

FIG. 9B depicts the key waveforms of the circuit from FIG. 9A.

FIG. 9C1 presents the formula of the parasitic energy in two transistorsforward topology.

FIG. 9C2 presents the formula for the peak negative magnetizing currentin the circuit from FIG. 9A.

FIG. 10A presents the two transistors forward topology using theintelligent clamp circuit.

FIG. 10B presents the key waveforms of the circuit from FIG. 10A.

FIG. 10C depicts the key waveforms of the circuit from FIG. 10A whereinthe intelligent clamp circuit is programed for ZVS.

FIG. 11A is presented the flyback derived, single ended asymmetricalhalf bridge.

FIG. 11B presents the key waveforms of the circuit from FIG. 11A.

FIG. 11C is presented the flyback derived, single ended asymmetricalhalf bridge operating in discontinuous mode.

FIG. 12A depicts the flyback derived, single ended asymmetrical halfbridge using the intelligent clamp circuit.

FIG. 12B depicts the flyback derived, single ended asymmetrical halfbridge using the intelligent clamp circuit.

FIG. 12C depicts the key waveforms from the circuit from FIG. 12B.

FIG. 13A depicts the flyback derived, single ended asymmetrical halfbridge using the intelligent clamp circuit placed on an auxiliarywinding.

DETAILED DESCRIPTION

In this specification, two modules are introduced. The first moduledepicted in FIG. 1B, 612 , is referred to as “Controlled EnergyPreservation Module”, CEPM. It has two power terminations A, 4000 and B,4010, and a signal connection, Cy, 4090.

The CEPM, 612 is composed by a unidirectional switch formed by diode D2,520, and a controlled switch Sy, 4020, and a controlled voltage source,Vvinjc, 4033. All these three said components are placed in series inany order. The presence of Vvinj,4033, allows an injection of energy fora given purpose into the power converter wherein the CEPM, 612, moduleis connected. The said purpose is to obtain zero voltage switching onthe switching elements.

The switch Sy, 4020, is controlled by the signal Vcsy, 4400 which isconnected to the signal connection Cy, 4090.

In FIG. 3D, FIG. 3E and FIG. 3F are depicted a possible implementationsof the circuit from FIG. 1B wherein the Vinjc, 4033 can be modulated.When Sy′ is closed the diode D2, 520 is turned off and the current fromA to B would flow via D3, 273, Vinj+,270, and Vvinj, 4030. In conclusionthe controlled voltage source Vinjc, 4033, from FIG. 1B, can have thevalue of Vvinjc as follows: when Sy′ is open Vvinjc=Vvinj, andVvinjc=(Vinj+Vinj+) when Sy′ is closed. Vinj can have a value as low aszero.

In FIG. 3F when Sy is closed the Vinjc is Vvinj and when Sy′ is closedand Vinj+ is larger than Vinj, the equivalent Vinjc is Vinj+. In FIG. 3Ethe implementation for Vinjc is a capacitor Cinjc charged by a currentsource Iinjc, 4013. In all these implementations the controlled voltagesource Vinjc can be modulated as needed for a given application.

In FIG. 1C is presented the Current Injection Module (CIM), 630 is amodule which in some application may work together with CEPM, toguaranty zero voltage switching conditions in any operating conditions.The CIM module is formed by a coupled winding to the transformer whereinthe CEPM module is attached and further containing a switching elementMinj, 550, a diode Dinj, 590, an optional Cinj capacitor, 570, whereinthe diode Dinj is connected to an external voltage source, Vinj. Thiscircuit is presented in the U.S. Pat. No. 10,574,148, wherein thisspecification is a continuation in part.

In FIG. 4A is presented the boost topology using the CEPM module. The Atermination of the EPM is connected to termination W1, 4110, of theinductive element L, 220, and the B termination of the CEPM module isconnected to the W2, 4100, termination of the inductor L, 220. Thecontrol signal Vcsy, 4400 is connected to the connection Cy of the CEPMmodule. In FIG. 4B are depicted the key waveforms of the circuitdepicted in FIG. 4A.

In between t0 to t1, the switch M1, is turned on by the signal VcM1,320, and during this time the current through L,220, builds up reachinga peak current Ipk,360. During this time interval, energy is accumulatedin the inductive element L. In this topology the first switch is M1, 210and the second switch is Do, 240 and the Sy switch is complementary tothe first switch.

At t1, the switch M1, 210, is turned off and the energy stored inbetween t0 to t1 is delivered to the output load. The current through L,220 will reach zero at t2, which means that all the energy stored in L,220, is delivered to the output load.

In between t2 to 3′, is the period wherein the unidirectional switchformed by Sy and D2 together with Vvinjc creates a low impedance pathacross the inductive element L, 220. The parasitic energy contained inthe parasitic oscillation is converted in magnetic energy viamagnetizing current I(Sy), 440. With Vvinjc of zero value, themagnetizing current would decay towards zero, as depicted by 4120, dueto the conduction losses. The value of Vvinjc will determine themagnetizing current shape during the t2 to t3′. The Vvinjc can bechosen, such as the amplitude of the magnetizing current through L,I_(L), 290 to be constant during t2 to t3′. The Vvinjc can be alsochosen such that the amplitude of the magnetizing current to increaseduring the period t2 to t3′ and in this way having the necessaryamplitude to discharge the voltage across Cp, 300 to zero between t3′ tot3. For a large “dead time” this implementation, maintaining constantamplitude I(Sy), 440, will lead to a larger power dissipation inconduction.

In FIG. 4C are depicted the key waveforms wherein intelligent control isused for driving the CEPM. The Vvinjc, 4033 can change and increase itsamplitude at a given time before t5 such as t4, as presented in FIG. 4C.By increasing the Vvinjc amplitude at a given time prior t5 themagnetizing current I_(L) would increase in a such way that between t4and t5 will reach the necessary amplitude to do a full discharge of theCp, 300, prior M1, 210 will turn on at t6 under zero voltage switchingconditions.

In FIG. 4C are presented the control signal for M1, VcM1, 320, thecontrol signal for Sy, Vcsy, 4400, the voltage across the main switchM1, V(M1), 4010, the current through Sy, I(Cy), 440, the current throughL, I_(L), 290, the Vvinjc, 4033.

In between t0 to t3 of FIG. 4C the mode of operation is the same as FIG.4B mode of operation between t0 to t2. In FIG. 4C, at t3 the currentthrough L, 220, reaches zero as is also presented in FIG. 4B at t2.

In FIG. 4C in between t3 to t4 the current through Sy, is presented asconstant, though it can be also decaying in the event Vvinjc has a verylow amplitude or even zero.

At t4 the voltage between A, 4000 and B, 4010 of the module CEPMincreases from Vinj, to an additional Vinj+, 270. This increase is alsovisible on the voltage across M1, during t4 to t5 and as a result thecurrent through Sy, starts increasing with a slope proportionate withthe Vvinj+, 270.

The increased amplitude of the current through Sy, increases also thenegative magnetizing current through I(L), 290 between t4 to t5.

At t5, the negative current through L, 220, has the necessary amplitudeto discharge the parasitic capacitance Cp, 300 to zero after t5, whenSy, 4020, is turned off. As a result, the voltage across M1, 210, willbe zero at t6, when M1, 210, turns on.

Using the CEPM module depicted in FIG. 4G, at t4, Sy′ is turned on viathe signal connection Cy′ which turn on Sy′, 4030 at t4 and turns it offat t5.

In this CEPM module depicted in FIG. 3D the Vvinjc will have a variableamplitude ranging from Vvinj to Vvinj+Vvinj+. Modulating the equivalentVvinjc will control the value of the current through Sy, 4020 and as aresult the current through I_(L), 290 to have the necessary amplitude att5, from to discharge the parasitic capacitance across Cp, 300 to adesired value. In most of the applications the desired voltage valueacross Cp, is zero voltage at t6 as depicted in FIG. 4C.

The modulation of the equivalent Vvinj can be done in many other waysdifferent of the implementation from FIG. 3D such as FIG. 3F and FIG.3E.

In FIG. 1A there are two switchers, Si, 110 and Sn, 130. We introducethe concept of the first and second switch. The second switch is theswitch which after turn off the parasitic oscillations start to occur.The first switch is the switch which is on prior the second switch andin between the first switch and second switch there is a small deadtime. A small dead time is much shorter than the dead time whereinparasitic oscillations do occur.

For example, in the boost topology, the first switch is M1 and thesecond switch is Do.

Parasitic oscillations do occur also to other topologies such as BuckTopology. In FIG. 5A is presented the buck topology and in FIG. 5B arepresented the key waveforms associated with the buck topology from FIG.5A.

The key waveforms depicted in FIG. 5B are: the control signal for M1,VcM1, 1050, the current through M1, IM1, 1100, the current through Lo,I(Lo), 1110, and the voltage in the switching node A, V(A), 1120.

Between t0 to t1 the switch M1, 1030, turns on and during this timeinterval the current builds up through M1, from zero to a peak current.As visible in FIG. 5B there is a spike of current which occurs at theturn on of M1, due to the discharge of the parasitic capacitance Cp1,302, reflected across M1, 1030.

Between t1 to t2 the current through Lo, 1070, decreases with a slopeproportionate to Vo/Lo. At t2 the current through Lo,1070, reaches zerowhich means that the energy injected in the magnetizing current of Lo isfully transferred to the load. I am defining as a load the outputconfiguration connected to Vo, composed by the output capacitor Co, 333and output load, 330.

In this topology the first switch is M1 and the second switch is M2.

The time in between t2 to t3 is referred as the dead time, time whereinneither of the switching elements, M1, 1030 and M2, 1040 is on. Duringthe dead time the voltage in A,280 exhibits parasitic oscillation causedby the resonant circuit formed by the inductive element, Lo, 1070 andthe parasitic capacitors across M1 and M2 which are in parallel. Theparasitic oscillations are energized by W_(pBuck), which is depicted inFIG. 5C.

In FIG. 5D is presented the buck topology using the module CEPM, and thekey waveforms are depicted in FIG. 5E. The key waveforms depicted inFIG. 5E are: Control signal for M1, Vc_(M1), 1050, the control signalfor the Sy, Vc_(Sy), the current through the main switch M1, thecurrent, I_(M1), 1100, the current through output inductor, I(Lo), 1110,the current through the switch Sy, 222, and the voltage in the switchingnode A, V(A), 280.

Between t0 to t1, M1, 1030 is on and the current build up through Lo,1070. At t1 M1 turns off and the current flows further through Lo,through M2, decaying and reaching zero level at t3.

At t2 in between t1 to t3 the Sy switch turns on via the Vc(Sy),1055. Aspreviously mentioned, the switch element of the module CEPM, (Sy) iscomplementary to the first switch (M1).

At t3 when the current through Lo reaches zero, the parasiticoscillation would start without the presence of the CEPM module.

In between t3 to t4 the voltage in the switching node A, will reach thelevel Vo-Vvinjc. Initially the energy WpBulk, presented in FIG. 5C willconvert in magnetizing current, Irr, of Lo which will flow through Sy,during the interval, t3 to t4. In some applications Vvinjc will ensurethat the amplitude Irr, will be maintained in between t3 to t4, thevoltage across M1. The amplitude of Vvinjc can be tailored after t4, toreach a higher amplitude for the purpose to increase the negativemagnetizing current of Lo, in such way to be able to discharge theparasitic capacitance reflected across M1, to zero shortly after t5,when the main switch M1 turns on. In between t4 to t5 Vcinjc willincrease its amplitude and that is reflected in V(A), 280.

The time interval between t4 to t5 and the amplitude of Vvinjc will betailored in such way that magnetizing current will be sufficient tofully discharge the parasitic capacitance reflected across M1 andachieve zero voltage switching conditions at the time M1 turns on. Thedesigners has two parameters to control in order to obtain zero voltageswitching on M1, one being the amplitude of Vvinjc and the time intervalbetween t4 to t5. In most of applications Vvinjc is constant and thetime difference between t4 to t5 can be easily controlled digitally.

Parasitic oscillations do occur also in other topologies such asisolated topologies including the flyback topology. In FIG. 6A ispresented an isolated flyback topology. An isolated flyback topology hasa primary and a secondary. In the primary section there is an inputvoltage source, Vin, 230, a transformer Tr1, 1200 having a primarywinding Lp, 1210 and a secondary winding Ls, 1220, and a primary switchM1, 210. The secondary circuit contains a secondary switch SR1, 1040 andthe output load composed by an output capacitor Co, 250 in parallel withan output load, R_(Load), 330.

The key waveforms of the flyback topology presented in FIG. 6A, aredepicted in FIG. 6B.

In between t0 to t1 the primary switch M1, 210 is turned on and thecurrent builds up through the primary winding storing energy in themagnetizing inductance of the transformer Tr1, 1200.

At t1 the primary switch M1, turns off and the magnetizing current willreflect from the primary winding to the secondary winding via switchSR1.

At t2 the current through the secondary winding reaches zero, whichmeans that the energy stored in the transformer Tr1, 1200 during t0 tot1 was fully delivered to the secondary between t1 to t2.

The time interval between t2 to t3, wherein primary and secondaryswitchers are off is the dead time. During this period parasiticoscillations do occur in the switching node, A, 280. These parasiticoscillations are caused by the resonant circuit formed by the primarywinding, 1210 and the parasitic capacitance reflected across M1, Cp,300, and energized by the Wp_(Flyback), depicted in FIG. 6C, wherein Nis the turn ratio between primary and secondary.

In the topology the first switch of the circuit from FIG. 1A is M1, 210and the second switch (Sn, 130) is SR1. As previously mentioned, theparasitic oscillation will start after the secondary switch turns off,in this topology after SR1, 1040, turns off.

In FIG. 7A is presented a flyback topology wherein CEPM module is addedacross an auxiliary winding Lx,450. The CEPM can be also be placedacross the primary winding Lp,1210, though in this case a floating drivefor Sy, 1020 of CEPM will be necessary. The module is the CEPM, 612. Itis formed by the switch Sy, 1020, diode D2, 520 and the controlledvoltage source CEPM. The role of this module is to extract theelectrical energy contained in the parasitic capacitor of Cp, 300, andconvert it in the magnetizing current in the Tr1, 1200 and in additionto that inject a controlled energy in magnetizing current of Tr1, 1200by the controlled voltage source Vvinjc of CEPM. There is a controller,4444, which controls M1, M2, the switch Sy, 1020 via Cy, 4090, andcontrol of Vvinjc.

In FIG. 7B are depicted the key waveforms for the circuit from FIG. 7Awherein only CEPM module is used. In this topology the first switch isM1, and the second switch is SR1, 1040. The control for the Sy iscomplementary signal of the first switch and that is depicted in FIG.7B. There is a dead time between the first switch, controlled by VcM1and the control signal for the Sy, which is Vcsy, as depicted in FIG. 7Bwhich is the time between t1 to t2 and the time in between t6 to t7.

The waveforms which are presented in FIG. 7B are: the control signal forM1, Vc_(M1), the control signal for the Sy, 4090, which is Vcsy,500, thevoltage across M1, V(M1), 280, the magnetizing current through TR1,1200, and the current through the auxiliary winding Lx, 450.

In between t0 to t1 the main switch M1 is turned on and the magnetizingcurrent builds up to a peak level at t1. After t1 turns off themagnetizing current starts flowing in the secondary winding and startsdecaying towards zero, reaching zero amplitude at t3. In between t1 tot3 all the energy stored in the magnetizing current of Tr1, is deliveredto the output load formed by Co, 333 and the output load, R_(Load), 330.

At t3, the energy stored in the parasitic capacitance Cp, 300 reflectedacross M1 the energy is Wp_(Flyback), defined in FIG. 6C.

In between t3 to t4 the energy stored in Cp, 300 is converted fromelectrical energy to magnetic energy, stored in the magnetizing currentIr, 4021 from FIG. 7B. The magnetizing current IM(Tr1) is flowing in thewinding Lx, via the diode D2, 520, Sy, 1020 and Vvinjc, 4033. In caseVvinjc has zero voltage, the amplitude of the Ir current will decreasein time, as depicted by 4041, in FIG. 7B, due to the conduction lossesin D2 and Sy, 4020. In some applications the role of the voltageinjection Vvinjc, 4033, in between t4 to t5 is to compensate for theamplitude loss of the current Ir. One of the embodiments of thisspecification is to increase the value of Vvinjc between t5 to t6. Somesolutions are depicted in FIG. 3D, FIG. 3E and FIG. 3F. By increasingthe amplitude of Vinj c, the amplitude of Ir should increase in such away that after t6 when the Sy, 4020 turns off the necessary energy inIM(TR1) is available to discharge the parasitic capacitance Cp, 300 tozero before the main switch M1, 210 turns on at t7. Through differentmeans analog or digital means the Vvinjc can be modulated in a way thatthe amplitude if Ir at t6 is satisfactory to discharge Cp, 300 to zeroor to a given voltage level. These features are available in theControlled Energy Preservation Module (CEPM). By controlling the valueof the time interval between t5 to t6 and by controlling the amplitudeof Vinjc during this interval the peak negative magnetizing current ofTr1 can be controlled in order to discharge the Cp,300, to a desiredvoltage level, in between t6 to t7. In most of the applications thedesired level is zero voltage, or a small voltage level in between t4 tot5 to achieve maximum efficiency of the power converter using theFlyback topology incorporating the CEPM module.

In FIG. 8A is presented a flyback topology circuit wherein two auxiliarywindings, Lx, 450 and Ly, 580 are coupled together with Lp, 1210 and Ls,1220. The Current Injection Module, CIM, is added to the Flyback circuittogether with the CEPM module. For the application wherein Vvinjc is notmodulated zero voltage switching is accomplish by using CIM, 630. InFIG. 8B are presented the key waveforms, VcM1, VcSy, V(M1), IM(Tr1),VcMinj, and the I(Ly).

Between t0 to t4 the mode of operation is exactly as the implementationfrom FIG. 8B which waveforms are depicted in FIG. 7B. In between t4 tot5, the magnetizing current Ir, 4021, represents the energy of theparasitic capacitance Cp, 300 converted in magnetizing current, Ir,between t3 to t4. The energy contained in the magnetizing current Irr,may not be enough to discharge the parasitic capacitor Cp, 300 to zeroafter t5 and before t6. In order to obtain zero voltage switchingconditions for M1 at t6, when M1 is turned on, at t5 the currentinjection circuit is activated, and a pulse of current will startflowing through the current injection winding, Ly, 580, current whichwill reflect into the primary winding Lp, 1210, and discharge theparasitic capacitance Cp, 300 towards zero prior t6 in such way thatwhen the switch M1, 210 turns on at zero voltage switching conditions.The discharge of the parasitic capacitance Cp, 300, is done by thenegative magnetizing current Ir, and by the current injection reflectedinto the primary winding and originated in the Ly winding. The currentinjection switch can be activated after the voltage across the mainswitch is decayed to a certain level by the magnetizing current in orderto optimize the discharge of the parasitic capacitance Cp, 300 to zero.The Vvinjc can be zero or a low amplitude in order to decrease theconduction power dissipation caused by Irr during the dead time. Byusing both modules, CEPM and CIM can create a flyback topology withoutparasitic oscillations during the dead time, Flyback topology which canoperate at constant frequency. The Vinj, 600 can be fully or partiallyenergized by other parasitic energies such as leakage inductance. Thissolution eliminates the need for valley detection and allows anoperation at higher frequency without jumping from a valley to anothervalley which leads to challenges in control.

This solution has many advantages. For example, for large dead time, themagnetizing current is decayed, and the current injection will alwaysguarantee zero voltage switching.

Another advantage is the capability to operate a constant frequency orchange the frequency of operation at as needed for the optimization ofthe system wherein this power converters is powering. The self-adjustingfeature of the Rompower current injection in the CIM wherein theamplitude of current injection is decreasing when the voltage across themain switch is decreasing, these two modules operate very efficientlytogether obtaining the best power conversion efficiencies by comparisonwith other solutions.

Another isolated topology wherein parasitic oscillations do occur is theTwo Transistor Forward Topology. In FIG. 9A is depicted the basiccircuit of a two-transistor forward topology.

The Two Transistor Forward Topology presented in FIG. 9A is composed bytwo primary switching devices in the primary, M1, 2030 and M2, 2040,controlled by two control signals VcM1, 2031 and VcM2, 2041. Atransformer Tr1, 2282 having a primary winding L1, 2010 and a secondarywinding L2, 2020, wherein the primary winding L1, 2010 is connected tosaid primary switching devices, is also part of the circuit. Dr1, 2111and Dr2, 2112 are the reset diodes and are placed across the primarywinding. In the secondary there are two synchronous rectifiers, SR1,2030 and SR2, 2061, an output inductor, Lo, 2070 connected to the outputload circuit, composed by an output capacitor Co, 333 and a R_(Load),300.

In FIG. 9B are presented the key waveforms of the topology presented inFIG. 9A. The key waveforms presented in FIG. 9B are: VcM1 & VcM2, thevoltage across M2, V(M2), 2200, the current through the primaryswitching element, I(M2), 2210 and the magnetizing current through TR1,IM(Tr1).

Between t0 to t1 the primary switchers M1, 2030 and M2, 2040 are on andcurrent starts flowing through said switchers and the primary winding ofthe transformer, L1, 2010. In the secondary the current flows throughSR1, 2030, and Lo, 2070 towards the output load formed by Co, 333 andR_(Load), 330. During this time period the magnetizing current alsobuilds up as depicted by IM(Tr1), via the current through the primarywinding L1, 2010, which represents the summation of the currentreflected from the secondary and the magnetizing current.

Between t1 to t2 the magnetizing current will open diodes Dr1 and Dr2and during this time the reset of the transformer Tr1, 2282 will occur.

At t2 the parasitic capacitances reflected across M1 and M2, have storedan energy which is presented in FIG. 9C1.

In Between t2 to t3, the energy stored in the parasitic capacitances Cp1and Cp2 will convert in magnetic energy via the magnetizing current IMx,2070 whose formula is presented in FIG. 9C2.

In between t3 to t4, also named the dead time period wherein parasiticoscillation do occur, oscillation created by the resonant circuit formedby the inductance of the primary winding and the parasitic capacitancesreflected across the switching elements, Cp1 and Cp2. This resonantcircuit is energized by the electrical energy defined by the formulaWp-TT forward presented in FIG. 9C1. The electrical energy defined inFIG. 9C1 is converted in magnetic energy during the time interval t2 tot3, magnetic energy stored in the transformer Tr1, 2282, by themagnetizing current IMx, 2070 defined in FIG. 9C2.

In FIG. 10A is presented the two-transistor forward circuit from FIG. 9Awherein the module, CEPM is placed across an auxiliary winding Lx, 450.

In FIG. 10B are depicted the key waveforms of the circuit presented inFIG. 10A.

In between t0 to t1 both switchers M1,2030 and M2, 2040, are turned onand during this time the magnetizing current will build up reaching itspeak at t1.

In this topology the primary switchers M1 and M2 form the first switch.The diodes Dr1 and Dr2 form the second switch. The parasitic oscillationstart after the second switch (Dr1&Dr2) turn off as depicted in FIG. 9B.The switch Sy 1020 which is part of the CEPM is complementary to thefirst switch with a small dead time in between, dead time, created bythe time interval between t1 to t2 and t5 to t6. This switch is alsocalled the “shorting switch”. The small dead time is called primaryswitch-shorting switch dead time. This is to differentiate it from thedead time in between the falling edge of the second switch and therising edge of the primary switch, dead time wherein the parasiticoscillations do occur. In FIG. 10B the on time for the second switch isthe time interval between t1 to t3 when the current through primarywinding, L1, 2010, reaches zero.

At t2, the switch Sy, 1020 of the CEPM is turned on.

In between t1 to t3 the magnetizing current decreases to reach zero att3. The time interval t1 to t3 is referred to as the reset time of thetransformer Tr, 2282. The energy stored in the leakage inductance inbetween the primary winding L1, 2010, and secondary winding, 2020, istransferred back to the input voltage source Vin, 230, after t1 when theswitchers M1 and M2 are turned off.

At t3, the parasitic capacitances reflected across M1 and M2, have anenergy W_(p-TT forward), whose formula is presented in FIG. 9C1. Inbetween t3 to t4, electrical energy stored in the parasitic capacitancesC1 and C2 is converted into magnetic energy via the magnetizing currentIMx, 2070.

At t4 the magnetizing current IM(TR1) reaches a negative amplitude IMx,2070. In between t4 to t5 the magnetizing current IMx which flows viaD2, 520, Sy, 4020, and Vvinjc, 4033, starts decaying due to theconduction losses as depicted by 2301 in the event wherein the Vinjcdoes not have the necessary voltage level to keep its amplitudeconstant. The necessary voltage level for Vinjc is the voltage levelwhich maintains a constant amplitude as depicted by 2305.

The voltage injection Vvinjc, 4033, can be tailored to achieve certaingoals for IMx. One goal is to maintain the necessary voltage level aspreviously mentions to maintain the amplitude of the IMx, constant asdepicted by 2305, in FIG. 10B. In such case, after t5 when Sy, 1020turns off at t5 the negative magnetizing current IMx, 2070, willdischarge the parasitic capacitances Cp1 and Cp2 to the voltage levelsof the first valley of the parasitic ringing. Other goal is to increasethe magnetizing current IMx to an amplitude such as at t5 themagnetizing current shall have the necessary amplitude to fullydischarge the parasitic capacitances Cp1, 2300 and Cp2, 2301 in such waythat the switchers M1, 2030 and M2, 2040 shall turn on at t6 at zerovoltage switching condition.

In one of the preferred embodiments of this specification, the voltageinjection source Vvinjc, 4033, shall have an increased amplitude for atime period prior to t5. This method has the advantage that it decreasesthe conduction losses by the IMx during the dead time, when IMx has alower amplitude and its amplitude is adjuated for a short time intervalbetween t5′ to t5. This concept is depicted in FIG. 10C. Usingintelligent control, the increase magnitude of Vvinjc shall occur at agiven time prior t5, named “T-ahead”, 2420. This method is the preferredsolution for maximum efficiency. The value of “T-ahead” and the V2amplitude between t5′ to t5, will control the amplitude of IMx at the t5as depicted by IM(TR1), 2440.

For example, in FIG. 10C the Vvinjc can be low amplitude, V1, 2422, fromt4 until t5′ and at t5′ Vvinjc shall increase to V2, 2424, amplitude.The “T-ahead” can be the result of a look up table in the intelligentcontrol. The delay in between t1 to t5′, after which Vvinjc swings fromV1 to V2, can be predicted in order for the magnetizing current IMx tohave the necessary amplitude at t5. The “T-ahead” can be easilymodulated digitally for a given V2, in order to achieve the desiredamplitude for the magnetizing current IM(Tr1) at t5.

This preferred embodiment is depicted in FIG. 10C wherein the amplitudeof Vvinjc can be small, V1, 2422, until t5′. At t5′ the amplitudeincreases from V1, 2410 to V2, 2430, which can be implemented by using aswitch as presented in FIGS. 3D and 3F. In case V2 has a determinedvoltage and the control of the negative magnetizing current 2440 is doneby tailoring the value of T-ahead.

This embodiment can apply to any topology wherein CEPM module isutilized, such as boost, buck, flyback, half bridge and half bridgederivative, full bridge and full bridge derivative and so on.

FIG. 11A illustrates a simplified schematic of electronic circuitry of apower converter utilizing the flyback derived, single ended asymmetricalhalf bridge topology, which is a half bridge derivative, wherein theembodiments of this specification can be applied. It is composed by atotem pole switchers, M1,3030, and M2,3040, and a transformer Tr1, 3282,which contains a primary winding L1, 3010 and a secondary winding, L23020 wherein the first termination of the primary winding is connectedto the common connection between M1 and M2, referred in thisspecification as switching node A, 119 and the second termination of theprimary winding is connected to the resonant capacitor C1, 3333. Thedrain of M1, 101 is connected to the input voltage source, Vin, 3230 andthe source of M2, 112 and the termination of C1, 3333 not connected tothe primary winding are connected to the negative polarity of the inputvoltage source Vin, 112.

The mode of operation of the single ended asymmetrical half bridge isdepicted in FIG. 11B. In FIG. 11B are depicted the key waveforms such asVcM1, 3031 and VcM2, 3041 which represent the control signal for M1,3030, and M2, 3040. In FIG. 11B is further presented the current throughL1, 5115 and the magnetizing current, 5120. Further in FIG. 11B ispresented also the voltage across the resonant capacitor C1, V(C1), 5116and the current through the secondary synchronous rectifier SR1, I(SR1),5117. Further in FIG. 11B is depicted also the voltage in the switchingnode, A, V(A), 5118.

The waveforms depicted in FIG. 11B apply for the operation in continuousmode. We define the continuous mode operation as the operation whereinthe Vc1M1 and Vc2M2 are successive to each other with a given dead timein between and without blanking phases. In discontinuous mode ofoperation there is an extended dead time following the on time of M2,102, when no energy is processed, said extended dead time which isseveral times larger than the said dead time between VcM1, 3031, andVcM2, 3041 in continuous mode of operation.

At lower output power the mode of operation is in discontinuous mode. Inthis mode the on time for M1 switch is followed by an on time of M2switch and followed by an extended dead time. In addition to themodulation of on time of the M1 switch the extended dead time can bealso modulated to decrease the power taken form the input. This mode ofoperation is depicted in FIG. 11C.

The mode of operation at very light load is by using a train of pulses,which are a succession of on time for M1 switch followed by on time forM2 switch, operation as described in continuous mode, followed by acontrol period of the extended dead time.

Like the flyback topology operating in discontinuous mode during theextended dead time there is an oscillation caused by the resonance inbetween the primary inductance L1, 3010, and the parasitic capacitancereflected in the switching node A, 119 as depicted in FIG. 11C.

In this topology M1 is the first switch and M2 is the second switch andthe inductive element depicted in FIG. 11A is the primary winding L1,3010 of the transformer Tr1, 3282.

As depicted in FIG. 11B between t0 to t1 the upper switch M1, 3030 isturned on and the current through the transformer primary winding, L1,3010 is building up until it reaches a determined peak level.

At t1 the upper switch M1, 3030, turns off and the magnetizing currentin the transformer Tr1, 3282 forces the conduction through the bodydiode of M2, 3040. The interval t1 to t2 by design is made to berelatively short to minimize the dissipation through the body diode.

At t2 the lower switch M2, 3040 is turned on and the magnetizing currentcontinues to flow through M2, L1 and resonant capacitor C1, 3333. Themagnetizing current is depicted in a dotted line, 5120. In addition tothe flow of the magnetizing current there is another quasi-resonantcurrent which is the result of the resonance in between the resonantcapacitor C1,3333 and the leakage inductance between L1, 3010 and L2,3020, of the transformer Tr1, 3282. The current reflected in thesecondary has a half sinusoidal shape. The half sinusoidal shape of thesecondary current, reflected in the primary via L1, 3010 is added to themagnetizing current flowing in the primary winding as depicted by I(L1),5115 from FIG. 11B.

At t3 the current in the secondary through SR1, 3030 reaches zero andturns off the SR1. The SR1 can be replaced by a diode function of theapplication. In this specification we will refer to SR1 as a“rectification means” which includes any rectification device whichconducts in one direction, and it is an open circuit when the currentreverses.

Between t3 to t4 the current in the primary winding L1, 3010, is reducedto the magnetizing current. The voltage across the C1 continues toincrease the magnetizing current into negative polarity. The longer thetime interval between t3 to t4 the larger the decay of the magnetizingcurrent.

Because the negative magnetizing current will charge the parasiticcapacitance reflected in the switching node A,119 and will flow furtherthrough the body diode of M1, 3030 creating zero voltage switchingcondition somewhere in between t4 to t5, the time interval t3 to t4 isan element in the design.

At t5 the upper switch M1, 3030, is turned on at zero voltage switchingconditions.

At t6 the magnetizing current, 5120, crosses zero and the cycle repeatsagain.

The operation in discontinuous mode depicted in FIG. 11C, hard switchingfor M1, 3030, may occur as depicted in FIG. 11C at t4. In FIG. 11C theM1,3030, switch is on in between t0 to t1, followed by a dead time fromt1 to t2 further followed by the turning on of M2 switch, between t2 tot3. During the “extended dead time”, which occurs after t3 there is anoscillation between the primary inductance L1, 3010 of the transformerTr1, 3282 and the parasitic capacitance reflected in the switching nodeA, 119. The parasitic capacitance reflected in the switching nodeincorporates the parasitic capacitances of M1, 3030, and M2, 3040, andadditional parasitic capacitance reflected across the primary winding ofthe transformer Tr1, 3282.

One of the embodiments of this specification is using the CEPM module,612 and place CEPM module across M2, as depicted in FIG. 12A. TheControlled Energy Preservation Module, CEPM, 612, is composed by aunidirectional switch formed by diode D2, 520, a switch Sy, 1020 and acontrolled voltage injection voltage source, Vvinjc, 4033. The CEPMmodule should be placed in between the switching node A and the sourceof M2, 3040, which is the input ground, GNDp, 112.

A preferred configuration is depicted in FIG. 12B. In the implementationof the FIG. 12B the voltage stress on the components in the CEPM isreduced.

In FIG. 12C are presented the key waveforms associated with the circuitpresented in FIG. 12A and FIG. 12B.

As presented in FIG. 12C in between t0 to t1 the upper switch M1, 3030is turned on. During this time period the current builds up via theprimary winding L1, 3010, the current in L1, 5115 during t0 to t1represents the magnetizing current through the transformer Tr1, 3282. Inbetween t0 to t1 energy is transferred from Vin, 3230 to the magnetizingcurrent of the transformer and in charging the capacitor C1.

At t1, M1, 3030 turns off and the current flowing through L1, 3010 willstart flowing through M2, initially through the body diode and at t2,M2, 3040 is turned on.

As previously mentioned, the control signal for the control of Sy it iscomplementary to the control signal of the first switch, which in thistopology is M1, 3030 with a dead time in between, in this case the deadtime is the time interval between t1 to t2′ and the time intervalbetween t5 to t6.

During t2 to t3 a quasi-resonant current pulse is flowing from theprimary winding to the secondary winding, quasi-resonant currentproduced by the resonant circuit formed by the leakage inductance inbetween L1 and L2, and the capacitor C1, 3333. Via said quasi-resonantcurrent, a quantum of energy is transferred from primary to secondary.The magnetizing current 5120 continues to flow into the primary windingfurther until t4, when the magnetizing current has a negative polarity,IMy.

Sy switch, 1020 is turned on sometimes between t2 to t4 at t2′.

At t4 the lower switch M2, 3040 will turn off. In the circuit from FIG.11A parasitic oscillations would occur in the switching node A, 119 andalso reflected in the magnetizing current as is depicted in FIG. 11C.The CEPM module from FIG. 12A creates a low impedance path via D2 and Syand Vvinjc of the CEPM. The magnetizing current IMy will start flowingvia D2, Sy and Vvinjc and the energy contained on the parasiticoscillations is converted in the magnetic energy via the magnetizingcurrent IMy. If the value of the Vvinjc is low or zero, the IMy woulddecrease in amplitude due to the conduction losses, as depicted by thedotted line, 5533. The value of Vvinjc can be chosen by differentcriteria function of the designer goals. For example, the value of theVvinjc can be chosen to keep the amplitude of the magnetizing currentIMy constant during the extended dead time. Another criteria is tomodulate the value of Vvinj in such way that at the end of the extendeddead time the amplitude of IMy is sufficient to discharge the parasiticcapacitance reflected in the switching nod A, to create zero voltageswitching conditions for M1, at t6 when the upper switch M1, 3030 turnson.

In FIG. 12C is presented such a concept wherein the Vvinjc which has aconstant amplitude. To minimize the conduction losses during theextended dead time, the amplitude of the IMx has to be kept low andprior t5, the amplitude of the negative magnetizing current IMy can beincreased by increasing the amplitude of the Vvinjc.

In FIG. 12C are depicted the key waveforms of the circuit presented inFIG. 12A and FIG. 12B. In FIG. 12C the value of Vvinjc is V1 between t0to t5′, and is chosen in such way that the amplitude of the magnetizingcurrent IMy to be constant.

At t5′ the amplitude of Vvinjc changes from V1 to V2 and the magnetizingcurrent amplitude increases from IMy to IMy′ which has the necessaryamplitude that at t5 to discharge the parasitic capacitance reflected inA, 119, and create zero voltage switching conditions for M2 at t6. InFIG. 3D, 3E, 3F are presented several solutions wherein the amplitude ofVvinjc can be modulated.

In FIG. 13A is presented another method of connecting the CEPM modulevia an auxiliary winding Lx which is tightly coupled with L1 and L2 onthe transformer Tr1. This mode of connection has the advantage of usinglower voltage components as part of the module CEPM and easier to drivethe Sy, 1020 wherein silicon mosfets or GaN or SiC devices are used.

In this topology the IMy can be tailored to have a certain value in suchway that if Vvinc is chosen to keep the amplitude constant, at t5 theamplitude of the magnetizing current IMy′ is sufficient to obtain zerovoltage switching for M1 at t6.

In another embodiment of this specification, we can combine the CEPMmodule with CIM like in flyback topology. In such case the Vvinjc can besmall or even zero and the current injection pulse will be initiatedprior the turn on of M1 at t6. The combination of CEPM and CIM will workvery well due to the amplitude self-adjusting feature of the CIM, and insome applications Vinjc of the CEPM can be zero for simplicity.

The CIM module can operate also in continuous mode in applicationwherein IMy will not have the necessary amplitude for turning on M1 atzero voltage switching.

In conclusion the two modules, CEPM and CIM can operate independently ortogether in all the topologies, isolated and non-isolated and ensure thezero voltage operation in any operating conditions.

Having a low amplitude Vvinjc such as maintain negative magnetizingcurrent constant and Current injection using CIM can ensure a veryefficiency operation in this topology and any other topology which hasdead time with parasitic oscillations.

What is claimed is:
 1. A switching power conversion apparatus, forconverting power from an input voltage source to a load, comprising: afirst switch connected to a switching node; a second switch connected tothe switching node, wherein the second switch is turned on after thefirst switch turns off; an inductive element having a magnetizingcurrent connected to the switching node, wherein the inductive elementis connected to deliver energy via the first and second switches fromthe input voltage to the load during a succession of power conversioncycles; a capacitance connected to the switching node wherein thecapacitance is configured to resonate with the inductive element duringa portion of the power conversion cycles to cause a parasiticoscillation unrelated to the power conversion cycles; a clamp subcircuitplaced across the inductive element containing an auxiliary switch totrap energy in the inductive element and prevent the parasiticoscillation, wherein the auxiliary switch is complementary to the firstswitch; and a controlled voltage source, controlled in an amplitude anda duration of the amplitude, to inject energy in the inductive elementto control an amplitude of the magnetizing current of the inductiveelement at a determined level, at a time when the auxiliary switch turnsoff to discharge the parasitic capacitance to a given voltage by usingtrapped energy in the inductive element in addition to injected energyfrom the controlled voltage source.
 2. The apparatus of claim 1, whereinthe inductive element comprises a choke.
 3. The apparatus of claim 2,wherein the power conversion apparatus is a non-isolated boost convertercontaining a power switch and a rectifier means, wherein the powerswitch is the first switch and the rectifier means is a secondaryswitch.
 4. The apparatus of claim 2, wherein the power conversionapparatus is a non-isolated buck converter containing a power switch anda rectifier means, wherein the power switch which is the first switchand the rectifier means is a secondary switch.
 5. The apparatus of claim1, wherein: the inductive element comprises a transformer, and thetransformer contains a primary winding, at least one secondary winding,and auxiliary windings; and the clamp subcircuit is placed across theprimary winding.
 6. The apparatus of claim 5, wherein the transformer ispart of a flyback converter.
 7. The apparatus of claim 5, wherein thetransformer is part of a forward converter.
 8. The apparatus of claim 5,wherein the transformer is part of a half bridge converter.
 9. Theapparatus of claim 5, wherein the transformer is part of a full bridgeconverter.
 10. The apparatus of claim 1 wherein the inductive elementcomprises a transformer containing a primary winding, at least onesecondary winding, and auxiliary windings, and the clamp subcircuit isplaced across the auxiliary winding.
 11. The apparatus of claim 10,wherein the transformer is part of a flyback converter.
 12. Theapparatus of claim 10, wherein the transformer is part of a forwardconverter.
 13. The apparatus of claim 10, wherein the transformer ispart of a half bridge converter.
 14. The apparatus of claim 10, whereinthe transformer is part of a full bridge converter.
 15. A switchingpower conversion apparatus, for converting power from an input voltagesource to a load, comprising: a first switch connected to a switchingnode; a second switch connected to the switching node wherein the secondswitch is turned on after the first switch turns off; an inductiveelement connected to the switching node, wherein the inductive elementis connected to deliver energy via the first and second switches fromthe input voltage to the load during a succession of power conversioncycles; the switching node is a primary winding of a transformer, thetransformer having at least a secondary winding and at least anauxiliary winding; a current injection circuit containing a currentinjection switch, a current injection diode, and a current injectionvoltage source connected across an auxiliary winding, wherein thecurrent injection switch is turned on prior to the first switch turningon and is turned off after the first switch is turned on; a currentinjection capacitor connected between a termination of the currentinjection switch which is not connected to the auxiliary winding and acathode of the current injection diode; a capacitance connected to theswitching node, wherein the capacitance is configured to resonate withthe inductive element during a portion of the power conversion cycles tocause a parasitic oscillation unrelated to the power conversion cycles;a clamp subcircuit placed across the primary winding containing anauxiliary switch to trap energy in the inductive element and prevent theparasitic oscillation, wherein the auxiliary switch is complementary tothe first switch; and a controlled voltage source to inject energy inthe inductive element to control amplitude of a magnetizing current ofthe inductive element at a determined level, when the auxiliary switchturns off to discharge the parasitic capacitance to a given voltage byusing trapped energy in the inductive element in addition to injectedenergy from the controlled voltage source.
 16. The apparatus of claim15, wherein the transformer is part of a flyback converter.
 17. Theapparatus of claim 15, wherein the transformer is part of a forwardconverter.
 18. The apparatus of claim 15, wherein the transformer ispart of a half bridge converter.
 19. The apparatus of claim 15, whereinthe transformer is part of a full bridge converter.
 20. A switchingpower conversion apparatus, for converting power from an input voltagesource to a load, comprising: a first switch connected to a switchingnode; a second switch connected to the switching node wherein the secondswitch is turned on after the first switch turns off; an inductiveelement connected to the switching node, wherein the inductive elementis connected to deliver energy via the first and second switches fromthe input voltage to the load during a succession of power conversioncycles; the switching node is a primary winding of a transformer, thetransformer having at least a secondary winding, and having first andsecond auxiliary windings; a current injection circuit containing acurrent injection switch, a current injection diode, and a currentinjection voltage source connected across the first auxiliary winding,wherein the current injection switch is turned on prior to the firstswitch turning on and is turned off after the first switch is turned on;a current injection capacitor connected between a termination of thecurrent injection switch which is not connected to the first auxiliarywinding and a cathode of the current injection diode; a capacitanceconnected to the switching node, wherein the capacitance is configuredto resonate with the inductive element during a portion of the powerconversion cycles to cause a parasitic oscillation unrelated to thepower conversion cycles, and a clamp subcircuit placed across the secondauxiliary winding containing an auxiliary switch to trap energy in theinductive element and prevent the parasitic oscillation; wherein theauxiliary switch is complementary to the first switch; and a controlledvoltage source to inject energy in the inductive element to controlamplitude of a magnetizing current of the inductive element at adetermined level, when the auxiliary switch turns off to discharge theparasitic capacitance to a given voltage by using trapped energy in theinductive element in addition to injected energy from said controlledvoltage source.
 21. The apparatus of claim 20, wherein the transformeris part of a flyback converter.
 22. The apparatus of claim 20, whereinthe transformer is part of a forward converter.
 23. The apparatus ofclaim 20, wherein the transformer is part of a half bridge converter.24. The apparatus of claim 20, wherein the transformer is part of a fullbridge converter.